An overload circuit is a protective topology that continuously monitors load current and disconnects power when it exceeds a safe threshold, preventing thermal damage to wires and components. Unlike a simple fuse that melts destructively, an active electronic overload circuit senses current in real-time and switches off a semiconductor device, allowing for instant, resettable protection.
In this guide, we will design a solid-state 12V DC overload protector capable of tripping at exactly 2.0A. We will cover the node topology, select exact component values, analyze extreme failure modes, and walk through a bench-testing procedure.
The Core Topology: High-Side P-MOSFET with Low-Side Shunt
To understand what an overload circuit is in practice, we must look at its signal chain. Our topology uses a high-side P-channel MOSFET to switch the load, controlled by a comparator that measures the voltage drop across a low-side shunt resistor.
Node Labels and Signal Path
- V_IN (Node 1): 12V DC source input.
- V_SENSE+ (Node 2): High side of the shunt resistor, connected to the load return.
- V_SENSE- (Node 3): Low side of the shunt resistor, tied to system GND.
- V_REF (Node 4): The precise trip-point voltage fed to the comparator's reference pin.
- V_OUT (Node 5): Switched power delivered to the load.
- GND (Node 6): Common system ground.
A traditional thermal breaker or fuse suffers from slow trip times (allowing let-through current to spike) and ambient temperature drift. A solid-state comparator circuit trips in microseconds. Furthermore, placing the shunt on the low-side (near GND) keeps the common-mode input voltage near 0V, which is ideal for standard comparators like the LM393 that cannot read signals near their positive supply rail.
Design Walkthrough: Sizing Real Components for a 2A Limit
Let us pick exact component values to build a 12V nominal (11.5V–13.8V automotive range) circuit that trips at 2.0A.
1. The Shunt Resistor (R_SHUNT)
We need a voltage drop that is large enough to measure accurately but small enough to avoid wasting power. We will target a 0.1V drop at 2.0A.
Using Ohm's Law: R = V / I = 0.1V / 2.0A = 0.05Ω.
Power dissipation: P = I² × R = 4 × 0.05 = 0.2W. To ensure the resistor does not drift in value due to self-heating, we will use a 0.05Ω 2W metal film resistor. The 10x power overhead provides excellent thermal stability.
2. The Reference Voltage (V_REF)
The comparator needs a rock-solid 0.1V reference. Deriving this directly from a noisy 12V battery line will cause erratic tripping. Instead, we drop the 12V to 5V using a 78L05 linear regulator, then use a voltage divider.
Using R1 = 47kΩ and R2 = 1kΩ:
V_REF = 5V × (1k / (47k + 1k)) = 0.104V.
This sets our actual trip point to 0.104V / 0.05Ω = 2.08A.
3. The Switch and Driver
We use an IRF9540N P-channel MOSFET for the high-side switch. It has an R_DS(on) of 0.117Ω, meaning at 2A it will dissipate roughly 0.46W—easily handled without a heatsink. The LM393 dual comparator features an open-collector output, which we wire directly to the MOSFET gate with a 10kΩ pull-up resistor to V_IN.
| Component | Part Number / Value | Purpose |
|---|---|---|
| U1 | LM393N (Dual Comparator) | Compares V_SENSE to V_REF |
| Q1 | IRF9540N (P-Channel MOSFET) | High-side load switch |
| U2 | 78L05 (5V LDO Regulator) | Stable voltage for V_REF divider |
| R_SHUNT | 0.05Ω, 2W Metal Film | Current sense element |
| R1, R2 | 47kΩ, 1kΩ (1% tolerance) | V_REF voltage divider |
| R_PULL | 10kΩ | MOSFET gate pull-up to V_IN |
Behavior Matrix and Extreme Failure Modes
Understanding what breaks at the extremes is critical for robust circuit design. Below is the behavior matrix showing how the circuit reacts to dynamic changes, followed by an analysis of catastrophic component failures.
| Condition | V_SENSE (Node 2) | LM393 Output | Q1 MOSFET State | Load Status |
|---|---|---|---|---|
| Normal Load (1.0A) | 0.05V | High-Z (Pull-up to 12V) | OFF (V_GS = 0V) | Powered |
| Overload (2.5A) | 0.125V | Sinks to GND | ON (V_GS = -12V) | Disconnected |
| Load Removed (0A) | 0.00V | High-Z | OFF | Powered (Standby) |
| V_REF Drifts Up | N/A | Remains High-Z longer | Stays ON longer | Trip point increases |
What Breaks at the Extremes?
If R_SHUNT shorts: V_SENSE drops to 0V regardless of current. The comparator never trips. If the load subsequently shorts, the IRF9540N will pass massive current until the silicon melts or the upstream power supply collapses.
If R_SHUNT opens: This is the most dangerous failure mode. If the shunt trace breaks while the load is connected, V_SENSE (Node 2) is pulled up to the full 12V supply through the load. The LM393's common-mode input range is V_CC - 1.5V. If V_CC is 12V, an input of 12V exceeds the absolute maximum rating, potentially causing phase reversal or destroying the input stage.
Always place a 1N4148 signal diode between V_SENSE (Node 2) and the 5V rail (anode to V_SENSE, cathode to 5V). If the shunt opens and V_SENSE spikes to 12V, the diode conducts, clamping the comparator input to a safe 5.7V and forcing the comparator to trip, safely shutting off the MOSFET.
Step-by-Step Breadboard Testing Procedure
Do not connect a sensitive microcontroller or expensive motor to your first prototype. Use a power resistor as a dummy load. Here is how to verify the design on the bench.
- Wire the Power Rails: Connect your bench power supply to 12.0V. Set the supply's current limit to 3.0A to protect your breadboard wires during faults.
- Establish the Reference: Wire the 78L05 and the R1/R2 divider. Use a multimeter to verify V_REF (Node 4) reads exactly 0.104V ± 2mV. If it is off, swap the 1kΩ resistor for a 1.1kΩ or 910Ω to tune it.
- Wire the Switch and Shunt: Connect the IRF9540N source to V_IN, drain to the load positive. Connect the load negative to V_SENSE+, and V_SENSE- to GND. Insert the 0.05Ω shunt between V_SENSE+ and V_SENSE-.
- Connect the Comparator: Wire LM393 V_CC to 12V and GND to GND. Connect V_SENSE to the inverting input (-) and V_REF to the non-inverting input (+). Wire the 10kΩ pull-up from the output to V_IN, and connect the output to the MOSFET gate.
- Test Normal Operation: Connect a 12Ω, 10W power resistor as the load (draws ~1A). The voltage across the shunt should read ~0.05V. The load should receive full power.
- Test the Trip Point: Swap the 12Ω resistor for a 4.7Ω, 20W resistor (draws ~2.5A). The shunt voltage will attempt to rise to 0.125V. The LM393 will immediately pull the gate low, cutting power. The load voltage should drop to near 0V, and the shunt voltage should collapse to 0V.
- Verify Hysteresis (Optional): Because the LM393 snaps off the load, V_SENSE drops to 0, which immediately makes V_SENSE < V_REF again, turning the MOSFET back on. This causes high-frequency oscillation (chatter). To fix this, add a 1MΩ feedback resistor from the LM393 output to the non-inverting input (+) to introduce 10-20mV of hysteresis, ensuring it latches off until the load is removed.
Frequently Asked Questions
What is an overload circuit in a motor starter compared to DC electronics?
In industrial motor control, an overload circuit typically refers to a bimetallic thermal overload relay (like the Schneider Electric TeSys LRD series). These rely on physical heat buildup to trip a mechanical contact, mimicking the thermal mass of the motor windings. In DC electronics, an overload circuit is solid-state, using shunt resistors and comparators to trip in microseconds based on exact electrical thresholds rather than thermal lag.
Why does my overload circuit trip immediately when I connect a capacitive load?
This is caused by inrush current. When you connect a load with large input capacitors (like a DC-DC converter or a motor controller), the empty capacitors act as a dead short for the first few milliseconds. The current spike easily exceeds your 2.0A threshold, tripping the protector before the capacitors charge. To fix this, you must add a soft-start circuit, use an NTC thermistor in series with the load, or implement a time-delay RC network on the comparator's reference pin to ignore the first 50ms of current.
Can I use an N-channel MOSFET instead of a P-channel for high-side overload protection?
Yes, but it requires a charge pump or a specialized high-side gate driver IC (like the TI DRV8701). An N-channel MOSFET on the high side requires the gate voltage to be driven 10V above the source pin. Since the source pin is sitting at 12V, your gate driver must output 22V. P-channel MOSFETs are universally preferred for simple, discrete high-side switching because they only require pulling the gate down to GND to turn on, which an open-collector comparator can do directly.
What is the difference between an overload circuit and a short-circuit protector?
While often used interchangeably in casual conversation, they protect against different fault profiles. An overload circuit protects against sustained currents that are slightly above normal (e.g., 120% to 200% of rated current) which cause gradual thermal degradation. A short-circuit protector must react to massive, instantaneous current spikes (e.g., 10x to 50x rated current) caused by a dead short. The circuit designed above acts as both, but in heavy industrial applications, magnetic trip elements handle short circuits while thermal elements handle overloads. For a deeper look at standard protection definitions, refer to the All About Circuits guide on overcurrent protection and the Texas Instruments LM393 datasheet for comparator specifications.
